Amplifier system with current-mode servo feedback

ABSTRACT

A system and method for compensating an amplifier apparatus for low frequency and/or DC components of an externally applied input signal as well as for any voltage offsets contributed by the amplifier circuitry. Band-limited servo feedback is applied to predetermined nodes in the forward gain path to null out unwanted signal components, leaving a residual signal that, when amplified, will be centered around ground, so that the full dynamic range of the amplifier system may be utilized. Consequently, the signal-to-noise ratio available at the output of the amplifier system will be maximized. The servo compensation may either operate in continuous time, or it may be held constant once a suitable level of compensation has been established, or it may be adjusted from time to time to accommodate slow variations of the average DC component of the input signal.

CROSS REFERENCE TO RELATED APPLICATIONS

This is a continuation in part of U.S. patent application Ser. No.10/831,885 filed on Apr. 26, 2004 and U.S. patent application Ser. No.10/443,230 filed on May 21, 2003 which are hereby incorporated byreference in their entirety.

FIELD OF THE INVENTION

The present invention relates to servo offset compensation for anamplifier, and more particularly, to controllable active feedbackcircuits which can substantially attenuate a DC signal component or apredetermined range of low frequency components present in the outputsignal of the amplifier.

BACKGROUND OF THE INVENTION

When it is desirable to use an amplifier to detect and amplify only thetime-varying components of a signal that may also contain a non-zerostatic or average DC component, it is necessary to remove or otherwisenull out this unwanted DC signal component. This may be achieved byinserting suitable high-pass networks in the forward signal path, or byapplying low frequency negative feedback around the amplifier to confera high-pass characteristic upon the overall amplifier system.

Overview of the Forward-Path Approach:

Provided that the frequencies of interest are well above DC (i.e., abovezero Hertz), coupling capacitors placed in the forward gain path of anamplifier may suffice to remove unwanted DC and low frequencycomponents. However, to avoid excessive amplitude and phase distortion,the effective cutoff frequency of each coupling capacitor operating inconjunction with the input impedance of the associated amplificationstage should be much lower than the lowest frequency of interest in theinput signal. If the required cutoff frequency is very low, physicallylarge and often expensive capacitors will be needed. In the special casewherein the input signal includes any square or rectangular waveformcomponents of relatively low frequencies (that is, as compared to thehigh-pass cutoff frequency), the shape of the waveforms will appeardistorted, exhibiting a ‘tilt’ in the waveform plateaus. This type ofdistortion can be troublesome in precision measurement applications,particularly if the shape of the waveform is of importance, mandatingthe use of even larger coupling capacitors. While it is possible toutilize compensatory algorithms (either in hardware or software) tocorrect such predictable frequency-related amplitude and phasedistortion, the preferred design practice is to avoid the distortion inthe first place.

In cases where the input is provided as a differential signal,capacitive coupling of the input signals becomes problematic, since twowell-matched capacitors will be needed to preserve adequate common moderejection. Unfortunately, obtaining matched capacitors is difficult atbest, and is often impossible if very low input frequencies must beaccommodated.

Generally, capacitors consist of conductive layers separated by layersof insulating material (e.g., dielectric), to form the equivalent of twogalvanically isolated parallel plates. All practical dielectrics exhibitsome degree of polarization when subjected to an electric field, and ittakes a finite amount of time and energy to change this polarizationwhen the applied electric field changes. This mechanism gives rise todielectric loss and dielectric absorption Accordingly, the non-idealbehavior of a coupling capacitor can distort the signal in a non-linearfashion, which modifications represent distortions in the signal. Indemanding applications (pro audio and precision metrology inparticular), such distortion, albeit small, is considered unacceptable.

Overview of the Negative Feedback Approach:

As is known, low frequency negative feedback may be applied around anamplifier to impart a predetermined high-pass characteristic to thetransfer function of the amplifier. A suitably configured passivefeedback network can be employed, comprising either low-pass elementssuch as inductors placed in series with the feedback signal to directlyblock high-frequency feedback signals, or capacitive shunt elementsdisposed to divert high-frequency components out of the feedback loop tolocal ground. If desired, a combination of both types of elements may beemployed provided that loop stability criteria are satisfied. However,when the signals-of-interest contain significant low frequency spectralcomponents, the size, cost and availability of the frequency determiningelements in the feedback loop can become problematic.

As an alternative to passive feedback, negative servo feedback may beemployed to significantly attenuate DC and low frequency signalcomponents in the main output of an amplifier system.

The term servo feedback implies the use of an active integrator within anegative feedback loop coupled from the output to the input of anamplifier system. In the most basic configuration, an integratorcomprises a high-gain differential amplifier provided with capacitivefeedback connected between its output and inverting input. Thenon-inverting input is coupled to a reference potential, while asampling resistor connects between the main output node and theinverting input of the main differential amplifier, so that theintegrator produces an output that represents the inverse of thetime-integral of the difference between the main amplifier's outputsignal and the reference potential. As is known, the DC gain of such anintegrator circuit will approach the open loop gain of the embeddeddifferential amplifier, and will fall off at 6 dB per octave as thefrequency increases.

The output signal from the integrator may be used directly as thenegative servo feedback signal, or it may control means that provide thenecessary feedback signal. Since the magnitude of the servo feedbackfalls off with increasing frequency, the effective closed loop gain ofthe amplifier system will increase with frequency until an asymptoticvalue is reached where the gain for AC signals will be equal to thenominal closed loop gain of the amplifier system, as determined in thenormal fashion. It is therefore seen that the application of negativeservo feedback around an amplifier system will impart a high-passcharacteristic to its overall transfer function.

In addition, since the servo loop operates to force the amplifier'soutput to maintain an average value of DC ground, the residual AC signalappearing at the amplifier's output will appear centered about the localcommon potential, thereby maximizing the useful dynamic range of theamplifier system.

An illustrative example of a conventional servo feedback design isdepicted in FIG. 1, wherein a high-gain differential amplifier,configured as an inverting servo integrator, is disposed to provide lowfrequency negative feedback around a main amplifier. Element 200,representing the primary forward path amplifier, is connected to providenon-inverting gain determined in the known manner by resistors 203 and204. Amplifier 400, selected to have large open loop gain, is operativeas a single-pole servo integrator due to the connections of localfeedback capacitor 409 and input resistor 403 disposed to sample themain output signal at node 601. Reference potential 501 is coupledthrough resistor 402 to non-inverting input 406 of amplifier 400. Theoutput signal of amplifier 400 constitutes negative servo feedback,conveyed through resistor 202 to non-inverting input 206 of amplifier200. Since the non-inverting input of amplifier 400 is connected throughresistor 402 to reference potential 501, the feedback loop will operateto force the output of amplifier 200 to maintain an average potentialequal to the reference potential, which is conventionally signal groundbut may be another predetermined DC potential if desired. Provided thatamplifier 400 has low input bias current, resistor 403 can assume a verylarge value without introducing significant offset errors, allowingcapacitor 409 to be small while still providing a relatively low cutofffrequency for the amplifier system.

Presumably, the input signal presented at node 101 is derived from a lowimpedance source, and the output of amplifier 400 also exhibits a lowimpedance. For the servo feedback to be effective, it must be subtractedfrom the input signal. This is accomplished by using resistors 201 and202 to form a voltage divider network, to create at its midpoint thealgebraic sum of the input signal and the servo feedback signal.Disadvantageously, this network is a voltage divider that generallyattenuates the input signal at all frequencies, degrading the overallsignal-to-noise ratio. Moreover, as a consequence of the high impedanceexhibited at node 206, any current flowing out of node 408 must flowinto the external signal source connected to input node 101, which isgenerally undesirable.

The latter problem can be overcome by adding unity-gain input buffer100, as shown in FIG. 2, however the input signal will still beattenuated by the voltage divider formed by resistors 201 and 202. Thevalue of resistor 202 could be increased in value to reduce theattenuation effect, but this would require a commensurately larger DCoffsetting signal from the output of amplifier 400, often requiring thatthe power supply voltages provided to amplifier 400 would also have tobe increased, inevitably adding to the overall design complexity andcost.

It is heretofore known to provide servo feedback as a current-modesignal. When current-mode feedback is provided by a high-impedancecurrent source, it may be injected into the forward gain path withoutthe use of a series resistor, thereby avoiding the attenuation problemdescribed above.

An illustrative example is provided in FIG. 3. In this design, theoutput of the servo integrator is coupled to the base of a transistor sothat its collector terminal provides the current-mode feedback signal.

The emitter of transistor 300 is connected to positive power supply 301,and receives a control signal coupled to its base node through resistor202 from output 408. To compensate for the inverting gain of transistor300, amplifier 400 must be configured as a non-inverting integrator.Provided that the input signal presented to input 101 has a negative DCbias potential, the servo loop will operate properly so that the averageDC value at output 601 will be essentially zero. As is known, a similarcircuit employing an NPN transistor could be arranged for input signalsexhibiting a positive DC bias. In either case, however, the feedbackcurrent is unipolar, so each circuit can only accommodate input signalshaving the correct polarity.

All of the circuits described hereinbefore are adapted to receive onlysingle-ended input signals. When common-mode noise may be a concern, itis preferable to use an amplifier having differential inputs.

The use of differential voltage-mode servo feedback adapted to provideoffsetting compensation for a differential amplifier system wasdisclosed in U.S. Pat. No. 6,411,098 B1 (hereinafter “the '098 patent”)and U.S. application No. US2003/0206021 by Laletin, and subsequently inU.S. Published application No. 2003/0206021, the entire contents of bothbeing incorporated herein by reference.

For convenience, preamplifier 30, shown in the original FIG. 1D of the'098 patent, is reproduced here as FIG. 4, wherein the original elementdesignators have been retained.

A pair of inputs, +Input and −Input, are connected to an externaldevice-under-test (henceforth, “DUT”), so that the potential across thedevice is conveyed to the non-inverting inputs of buffer amplifiers 104and 112; resistors 81 and 82 provide return paths for the input biascurrents of these buffer amplifiers. Primary amplifier 90 is aconventional differential amplifier, receiving inputs through resistors85 and 86. Amplifier 93, operative with resistor 91 and capacitor 92,constitutes a servo integrator referenced to ground at its non-invertinginput, whose output is conveyed through resistor 87 to the non-invertinginput of primary amplifier 90. Additionally, the output of servointegrator 93 is inverted by amplifier 96 and therefrom conveyed throughresistor 88 to the inverting input of amplifier 90 as a complementarynegative feedback signal.

Since the non-inverting input of servo integrator 93 is connected toground, the effect of these dual feedback loops is to force the averagevalue of the output of amplifier 90 to remain at ground potential, sothat only an amplified version of the AC components of the differentialinput signal appear centered about ground at main output node 32. Theforward gain of amplifier 90 is determined in the known manner accordingto the values of resistors 85, 86, 87 and 88, while the common moderejection ratio depends on the ratio match of resistor pairs 85/87 and86/88. Note that resistors 87 and 88 not only serve as gain settingelements, but are also the coupling elements that convey the servofeedback to the input of differential amplifier 90.

When this circuit is used in conjunction with a composite mixed-modebridge amplifier comprising both a current-mode driver (22) and avoltage-mode driver (24) as disclosed in FIG. 1D of the '098 patent,both an auto-polarity and a self-centering function may be realized. Thefeedback signal from the output of servo integrator 93 is used tocontrol a voltage-controlled voltage source (VCVS) driver connected tothat terminal of the DUT to which −Input node the overall preamplifiercircuit is connected. When a device exhibiting a DC bias is connectedbetween the input terminals of the preamplifier, the servo loop, inconjunction with the VCVS, will operate to establish an equilibriumpoint such that the potentials present at each of the input terminalswill be of opposite polarity. Furthermore, due to the balanced nature ofthe complementary servo loops, the magnitude of the potential at eachinput node will be precisely one half of the half of the total DC biasof the DUT, thus effectively “centering” the externally applied biasvoltage about local ground.

While this circuit design provides excellent performance as long asresistors 85, 86, 87 and 88 all have the same value, problems arise if again greater than unity is required of amplifier 90. To achieve highergains, the values of resistors 87 and 88 must be made proportionatelylarger, requiring that the servo feedback signals issuing fromamplifiers 93 and 96 increase in magnitude as well to maintain the sameamount of DC offsetting capability. This may require that amplifiers 93and 96 be provided with commensurately greater power supply rails, whichcan increase circuit complexity and cost.

SUMMARY OF THE INVENTION

The difficulties and shortcomings inherent in the designs describedabove are overcome by the inventive apparatus and method as set forthbelow.

In many applications where precision amplification of low-level signalsis required, it is desirable to remove certain unwanted components fromthe input signal prior to subsequent amplification. Such unwanted signalcomponents may exist in externally applied input signals or may arisewithin the amplification circuitry itself. The illustrative methods andsystems described herein according to the present invention are employedto eliminate unwanted DC and low frequency signals by providingcomplementary servo feedback without the need for blocking capacitors inthe forward gain path.

According to an illustrative embodiment of the inventive method,band-limited current-mode negative servo feedback is employed to confera high-pass output characteristic upon an amplifier apparatus, therebyachieving significant attenuation of DC and low-frequency signals.Consequently, the average DC value of the residual signal present at themain output of the amplifier apparatus is substantially equal to areference potential.

A servo feedback loop is controlled by a high-gain differentialamplifier configured as an integrator exhibiting a closed-loop gain thatdecreases with frequency at a rate of about 6 dB per octave. Theintegrator is provided at one input with a suitable reference potentialthat is usually ground, while the other input of the integrator isadapted to sample the main output signal. The integrator output iscoupled to at least one controllable current source, yielding acurrent-mode feedback signal that is conveyed to a suitable node in theforward gain path of the amplifier apparatus. The unwanted DCcomponents, or a limited range of low frequency signal components, arethereby substantially removed from the main output signal, leaving aresidual signal that exhibits an average value substantially equal to areference potential that is usually equal to ground. To maximize thedynamic range available in the forward signal path of the amplifierapparatus, no significant amount of signal amplification is appliedahead of the point where the servo feedback is injected into thecircuit.

One application of the inventive method relates to the measurement ofsmall time-varying signals that are present across the terminals of anelectrochemical accumulator that exhibits an intrinsic DC biaspotential. The accumulator may be either a single energy storage cell ora battery of cells. Significant information about a cell or battery canbe obtained from the characteristics of small time-varying voltagesdeveloped across the cell/battery in response to certain types ofcurrent excitation. In order to evaluate the condition or state ofcharge of the cell/battery, it is important to measure thecharacteristics of these small time-varying voltages with considerableaccuracy.

To overcome the limitations of the prior art and confer additionaladvantages, the present invention provides circuitry to apply, to anamplifier system, low frequency servo feedback proportional to thetime-integral of the output voltage of the amplifier system, in a mannerthat does not compromise other desirable operating characteristics.

An illustrative embodiment of the invention includes an integratoroperative in conjunction with a controllable current source having avery high output impedance which provides current-mode servo feedback toremove DC and low-frequency signal components from the output signal ofan amplifier system. Since the current-mode feedback signal comes from ahigh-impedance source, it may be conveyed directly to a high impedanceamplifier input node without significantly affecting the effectiveimpedance at these nodes.

According to an aspect relating to both single-ended or differentialinput amplifier apparatus configurations, the negative servo feedbackcauses the signals-of-interest (specifically, those residual componentsremaining in the input signal after the unwanted components are removed)to become centered about ground. Therefore, the signals may besubstantially amplified to take advantage of the entire dynamic range ofthe amplifier system.

In another illustrative embodiment, means are provided whereby a servocorrection value may first operate in a “track mode’ to establish anequilibrium condition, and a ‘hold mode’ to make the servo correctionsignal remain at a fixed value. The amplifier system can therebyaccurately amplify any subsequent variations in the input signal. Suchvariations may represent not only continuously time-varying AC signalsbut also step-wise variations representing changes in the DC signallevel.

In accordance with another illustrative embodiment, the basic servointegrator may be replaced with a generalized servo controller thatincorporates controllable signal processing functions in addition to atleast one servo integrator. Such functions may include, but are notlimited to, track and hold circuitry, a low-pass filter, a medianfilter, an envelope detector, a noise-discriminator, and a notch filterhaving an adjustable width, an adjustable depth of attenuation and anadjustable center frequency.

A single-ended embodiment of the present invention includes an amplifierdisposed to receive a single-ended input signal, along with means togenerate current-mode servo feedback that is conveyed into the forwardsignal path of the amplifier.

A first differential embodiment of the invention includes a singledifferential amplifier disposed to operate as the main forward pathamplifier, coupled with a pair of controlled current sources configuredto provide complementary servo feedback signals to the inputs of theamplifier. Additional unity-gain input buffer amplifiers may be added torealize a differential amplifier configuration that has high impedanceinputs.

A second differential embodiment provides both high gain and widebandwidth by employing a cascaded amplifier configuration commonlyreferred to as an instrumentation amplifier. A first gain stagecomprises a pair of cross-connected differential amplifiers to which areconveyed complementary current-mode servo feedback signals, and a secondgain stage comprises a single differential amplifier.

The present invention embodies certain novel features and improvements,as illustrated in the accompanying drawings and specifications, and asparticularly pointed out in the appended claims. However, it should beunderstood that changes could be made in various details withoutdeparting from the spirit and scope of the present invention.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing and other features and advantages of the present inventionwill be more fully understood from the following detailed description ofillustrative embodiments taken in conjunction with the accompanyingdrawings in which:

FIG. 1 is a schematic circuit diagram of a single-ended servo amplifierconfiguration as heretofore known in the art;

FIG. 2 is a schematic circuit diagram of a buffered single-ended servoamplifier configuration as heretofore known in the art;

FIG. 3 is a schematic circuit diagram of a single ended amplifierconfiguration with current mode servo feedback as heretofore known inthe art;

FIG. 4 is a schematic circuit diagram of a differential voltage-modeservo configuration as heretofore known in the art;

FIG. 5 is a schematic circuit diagram of a differential amplifierconfiguration with complementary current-mode servo feedback accordingto an illustrative embodiment of the present invention;

FIG. 6 is a schematic circuit diagram of a differential amplifierconfiguration with complementary current-mode servo feedback and havinginput buffers an an analog servo integrator according to an illustrativeembodiment of the present invention;

FIG. 7 is a schematic circuit diagram of a differential amplifierconfiguration with complementary current-mode servo feedback, inputbuffers and a digital servo integrator according to an illustrativeembodiment of the present invention;

FIG. 8 is a schematic circuit diagram of a differential amplifierconfiguration with cascaded stages and complementary current-mode servofeedback according to an illustrative embodiment of the presentinvention;

FIG. 9 is a schematic circuit diagram of an alternative differentialamplifier configuration with cascaded stages and complementarycurrent-mode servo feedback according to an illustrative embodiment ofthe present invention;

FIG. 10 is a schematic circuit diagram of a single-ended amplifierconfiguration with current-mode servo feedback according to anillustrative embodiment of the present invention;

FIG. 11 is a schematic circuit diagram of an alternative single-endedamplifier configuration with current-mode servo feedback according to anillustrative embodiment of the present invention; and

FIG. 12 is a schematic circuit diagram of another alternativesingle-ended amplifier configuration with current-mode servo feedbackaccording to an illustrative embodiment of the present invention.

DETAILED DESCRIPTION

An illustrative embodiment of an amplifier apparatus according to theinvention is depicted in FIG. 5. Differential amplifier 200, disposed inthe forward gain path of an amplifier apparatus, constitutes a primaryamplifier that receives a first input signal through impedance element201 at its non-inverting input 206, and receives a second input signalthrough impedance element 203 at its inverting input 207, to producemain output signal 601. In practice, the potential on a first terminalof an external device-under-test (henceforth, DUT) represents the firstinput signal, while the potential on its second terminal represents thesecond input signal.

Within this embodiment and for each of the embodiments to be describedsubsequently, impedance elements 201 and 203 are typically resistors;impedance elements 202 and 204 are generally resistors, but additionalcapacitors may be added in parallel with each of these to provide localfrequency compensation. The forward gain and common-mode rejection ratioof amplifier 200 are determined in the known manner by elements 201,202, 203 and 204.

Resistor 403 samples main output signal 601 and couples it to invertinginput 407 of amplifier 400, while resistor 402 conveys referencepotential 501 to non-inverting input 406. With the connection ofcapacitor 409 between inverting input 407 and output 408, a single-poleintegrator is formed having a time constant determined by capacitor 409and resistor 403. Output signal 408 represents a band-limited signalcorresponding to the inverse of the time-integral of the differencebetween the sampled signal and the reference potential. The gain of anintegrator achieves its maximum value at DC, and falls off at 6 dB peroctave with increasing frequency at 6 dB per octave, corresponding to alow-pass (e.g., bandwidth-limited) transfer function. When an integratoris used to control a negative feedback loop, it is conventionallyreferred to as a servo integrator.

Output signal 408 serves as the controlling input 706 (henceforth, the“CCS control signal”) for controlled current source 700 that provides adirectly proportional output current 708, which current may be either apositive (source) current or a negative (sink) current according to thepolarity of its controlling input signal. The symbol used for currentsource 700 should be henceforth understood to denote avoltage-controlled current source having bipolar (e.g., source/sink)output capability, a non-inverting transfer function and an outputimpedance typically exceeding at least several megohms. Additionally,signal 408 is coupled as the controlling input 756 to invertingvoltage-controlled current source 750, which provides an inverselyproportional bipolar output current 758. In similar fashion, the symbolused for current source 750 should be understood to denote avoltage-controlled current source having bipolar output capability, aninverting transfer function and an output impedance typically exceedingat least several megohms. These two current sources receive the same CCScontrolling input, and consequently they will provide output currents ofsubstantially identical magnitudes but of complementary (that is,opposite) polarities.

Signal 708, representing a first current-mode negative servo feedbacksignal, is conveyed from current source 700 to non-inverting input 206of primary amplifier 200, and signal 758, representing a secondcurrent-mode negative servo feedback signal is conveyed from currentsource 750 to inverting input 207. Due to the low-pass characteristic ofthis feedback signal, the transfer function of the amplifier apparatuswill also be band-limited, exhibiting a high-pass characteristic suchthat DC signals present in the differential input signal will experiencethe greatest attenuation, while the attenuation of AC signals will varyinversely with frequency.

When a device-under-test (DUT) exhibiting a DC bias across its terminalsis connected between inputs 101 and 151, the servo integrator willoperate to establish an equilibrium condition such that average DCpotential of the residual signal appearing at main output 601 will besubstantially equal to reference potential 501, conventionally ground(although a different potential could be used if desired). When thiscondition is achieved, the differential DC voltage present between nodes206 and 207 must also be substantially equal to zero. Since the inputimpedances, with respect to node 501 (ground), exhibited at input 101and input 151 are preferably the same, the potential of a biased DUTconnected to these inputs will appear “centered” about local ground,such that the magnitudes of the voltages at nodes 101 and 151 will beequal, but they will have opposite polarities.

No current can flow in or out of the high impedance inputs of amplifier200, or through resistors 202 and 204 because each of these resistorswill have zero volts across it. Accordingly, all of the servo feedbackcurrent must flow from current source 700, through resistor 201, throughthe DUT and then back through 203 to current source 750. Due to thecomplementary nature of the servo feedback currents, the voltagesdeveloped by these currents flowing through resistors 201 and 203 willbe equal and opposite. The sum of the voltages developed acrossresistors 201 and 203 must be equal in magnitude to, but opposite inpolarity from the average DC potential present across the DUT.Equivalently, the total voltage developed across these resistors by theservo feedback current precisely offsets the average DC potential of theDUT.

Having servo feedback current flow though the DUT may sometimes beundesirable, particularly if the DUT is an electrochemical device suchas an energy storage cell or battery. Such current will always flow outof the positive terminal of the DUT and back into its negative terminal,therefore appearing as a net discharging current. In an illustrativeapplication of the inventive method, the amplifier apparatus is employedto detect and amplify the voltage appearing across the terminals oflarge energy storage cells or batteries of cells. Provided that thesecells are sufficiently robust, they should be substantially immune tothe effects of small discharge currents, if the duration of suchdischarging is limited to short periods of time.

While the magnitude of this discharging current could be reduced byproportionately increasing the values of each of the four gain settingresistors 201-204, the noise performance and bandwidth of the circuitwill be compromised if these resistors are made too large. Therefore,this circuit may considered when differential voltage sensing isrequired, subject to the limitation that each DUT can be expected to besufficiently robust such that it can supply these small dischargecurrents without causing significant errors.

As mentioned hereinbefore, the forward gain of amplifier 200 isdetermined in the known manner by the relative values of resistors201,202, 203 and 204, while the common-mode rejection ratio (CMRR) isdetermined by the accuracy of the match between two ratios, specificallyof resistors 201/202 and resistors 203/204. Small matched capacitors maybe connected in parallel with elements 202 and 204 to providehigh-frequency compensation. Deviations from the design-center values ofthese several impedance elements will lead to inaccuracies in the gainand CMMR of the amplifier apparatus. Because the output impedance ofeach of the current sources is at least several decades larger than thetypical values used for resistors 201, 202, 203 and 204, it is expectedthat these servo feedback connections will have a minimal effect on theaccuracy of the differential gain and CMRR as determined by the criticalresistor ratios.

In certain specialized applications, the device-under-test (DUT) will bestimulated with an AC current provided by an external circuit, and theamplifier apparatus will be called upon to detect and measure theresulting AC voltage developed across the internal impedance of the DUT.In such applications, it is not uncommon for a significant ACcommon-mode signal to be present across the DUT, so having goodcommon-mode rejection capability becomes important.

When typical commercial components are used within the amplifierapparatus, the resulting CMRR may be insufficient for precisionmeasurements. Therefore, additional means may be provided to trim thevalues of the impedance elements surrounding amplifier 200 (i.e.,resistors 201, 202, 203 and 204) to maximize CMRR at DC and selected ACfrequencies. Moreover, although current sources 700 and 750 willtypically have large output impedances, these output impedances appearin parallel with the impedance networks at inputs 206 and 207,potentially leading to small gain and CMRR errors. Hence, controllablemeans may also be provided to adjust those output impedances whenexceptionally high accuracy is required. Such adjustment means may beincorporated as required in any of the embodiments of the inventivemethod.

An illustrative differential embodiment of the present invention isdepicted in FIG. 6, wherein the amplifier apparatus comprises a pair ofinput buffers, a primary differential amplifier and servo feedbackcircuitry. Complementary servo feedback currents are conveyed to theinputs of the primary differential amplifier to achieve DC andlow-frequency compensation in the main output signal.

The circuitry and operation of primary amplifier 200, integratoramplifier 400 and both controlled current sources remain unchanged fromthe previously described embodiment. The addition of the bufferamplifiers however, confers performance advantages as described below.

Input node 101 receives a first input signal from the first terminal ofthe DUT and unity-gain amplifier 100 provides a buffered copy at outputnode 108, while input node 151 receives a second input signal from thesecond terminal of a DUT and unity-gain amplifier 150 provides abuffered copy at output node 158. Due to the high input impedances ofamplifiers 100 and 150, the DUT is effectively isolated from the rest ofthe circuitry. Because the differential voltage present between nodes108 and 158 is substantially identical to the voltage across the DUT,the rest of the circuit will operate as described previously, with onesignificant difference: although the servo currents are stillconstrained to flow only through resistors 201 and 203, these currentsno longer flow through the DUT, but are absorbed instead by thelow-impedance outputs of unity gain amplifiers 100 and 150.

To ensure that the circuit functions properly in practical applicationswherein the DUT has no electrical connection to the amplifier apparatusother than through input connections 101 and 151, bias currents must besupplied to the non-inverting inputs of amplifiers 100 and 150. This maybe accomplished for each input amplifier by connecting its input througha very large valued resistor to local ground (or, optionally, anotherreference potential), recognizing that the values of these biasingresistors should be well-matched. Note that differential amplifiersconfigured in the conventional manner to operate as unity-gainnon-inverting amplifiers could replace the single-ended amplifiersdepicted in FIG. 6 without departing from the scope of the invention.

To further improve accuracy, the circuit can be augmented to includecircuitry to compensate for the input bias currents of each inputamplifier. Additionally, circuitry may be provided to adjust theresidual input voltage offset errors of each of the input amplifiers.

Another embodiment of the invention is depicted in FIG. 7. The analogservo integrator is replaced by a composite circuit implementedpartially in the digital domain, with suitable digital/analog conversiondevices in the input/output signal paths. A copy of output signal 601 isconveyed to ADC 802, where it is converted into a digitalrepresentation. An integrator algorithm, operative in suitable digitalhardware that may be a microprocessor or a Digital Signal Processor(DSP), performs numerical integration to yield an integrated digitalvalue implicitly referenced to the ground potential. This digital valueis conveyed to DAC 806 where it is converted into an analog controlsignal that is coupled to servo current sources 700 and 750. Thereafter,the amplifier and servo circuitry operates as previously described.

When the inventive amplifier apparatus is embedded within a larger testsystem that provides controllable excitation current through the deviceunder test to evoke a voltage response differential input capability,wide bandwidth extending to DC, substantial forward gain and accurateservo offsetting are all required. Such test systems include, but arenot limited to, electrochemical cell/battery testers and monitoringsystems, impedance testers and frequency response analyzers, time-domainresponse and spectrum analyzers, feedback-loop response analyzers,distortion analyzers, and biochemical test systems. The inventivecircuit is especially useful as an embedded sensing circuit, for examplein a battery charger, or a larger battery charging/monitoring system. Inthis application, the output of the inventive amplifier system isconveyed as a feedback signal to the system controller; this feedback isused to by the controller to evaluate the battery's condition, andthence, adjust the charging algorithm in real time to optimize chargingefficiency.

The condition and state of charge of an energy storage cell or batterymay be determined by analyzing the time-dependent polarization voltagethat appears across it in response to excitation by a suitabletime-varying current. Such analysis methods include Frequency ResponseAnalysis wherein sine wave excitation is employed, and Time DomainSpectroscopy based on square wave excitation.

For these techniques to yield the best information, it is important thatthe peak to peak amplitude of the polarization voltage response be keptsmall, not exceeding a few tens of millivolts. In some extreme cases thepeak-to-peak voltage may be on the order of one or two millivolts. It isdesirable to be able to measure this small voltage with an accuracy ofat least 1 part in 2000 (i.e., equivalent to 0.5% accuracy), soconsiderable amplification must be used in conjunction with circuitry tonull out the DC bias of the cell/battery. Additionally, to accuratelyreproduce the effects of square-wave current excitation, the frequencyresponse of the amplifier apparatus must extend all the way to DC. Theseperformance criteria are achieved by the design depicted in FIG. 8

To achieve the required amount of gain and still preserve wide-bandwidthcapability throughout the forward gain path, it is useful to employ anamplifier system that comprises two cascaded gain stages. A portion ofthe forward gain is provided by a first gain stage comprising a pair ofinput differential amplifiers that are cross-coupled in the known mannerto provide substantial differential mode gain but no more than unitycommon mode gain. A second gain stage comprises another differentialamplifier, representing the primary differential amplifier that receivesa differential input signal from the first stage, and outputs a mainsignal that is single-ended. This second gain stage is adapted toprovide at least unity gain for differential signals while significantlyattenuating common-mode input signals. In an embodiment of the inventivemethod, complementary current-mode servo feedback is provided to theinputs of the cross-coupled differential amplifiers, to null outunwanted low frequency components. By applying the servo feedback to thefirst gain stage, the DC signal components due to any intrinsic biasvoltage of a DUT are significantly attenuated before amplification isapplied at the first and second gains stages, thereby maximizing theuseful dynamic range throughout the forward gain path.

Differential amplifiers 100 and 150 comprise the first gain stage,accepting a differential input signal presented by a DUT connected toinputs 101 and 151, to yield an intermediate differential output signalcoupled to a second gain stage comprising primary differential amplifier200 and its associated components. Local negative feedback is conveyedto amplifiers 100 and 150 through equal-valued R_(FB) resistors 104 and154, respectively, while a single gain setting R_(G) resistor 135connects between the inverting inputs of the input amplifiers 100 and150. As is known, this cross-coupled arrangement ensures that each ofthe input amplifiers will have a common mode gain of only 1, while theoverall differential-mode gain of the amplifier pair will be{1+(2R_(FB)/R_(G))}. Small matched capacitors may be placed in parallelwith elements 104 and 154 to provide local high-frequency compensation.Controllable means may also be provided whereby the common moderejection of the first stage may be maximized both at DC and atpredetermined AC frequencies. By removing resistor 135, the inputamplifiers will each have a nominal gain of unity, whereupon thiscircuit becomes similar to the embodiment depicted in FIG. 7. But, sincethe cross-couple amplifier are now enclosed within the overall servofeedback loop, any intrinsic offset voltage errors attributable toamplifiers 100 and 150 will be substantially eliminated by the overallservo feedback.

Amplifier 200 constitutes the primary amplifier, representing the secondgain stage. Its input signals correspond to the intermediate outputsignals developed by the first gain stage. Amplifier 200 is configuredto reject common-mode signals while providing a gain equal to or greaterthan unity for differential mode signals, in a manner previouslydescribed. Controllable circuitry to adjust both the common moderejection ratio and differential gain of amplifier 200 may also beprovided. The output of amplifier 200 constitutes the main output signaland appears at node 601 of the cascaded amplifier apparatus, where it issampled by servo controller 90.

As shown in FIG. 8, servo controller 90 is adapted to provide controloutputs 91 and 92. In this embodiment these are analog control signalspreferably equal in magnitude but opposite in polarity, coupled tovoltage-controlled current sources 700 and 750, respectively.Alternatively, a single control signal output from servo controller 90could be used to drive both controllable current sources, provided thatone of the current sources has an inverting transfer function. It iswithin the scope of the invention to employ other types of controllablecurrent sources that may for example be digitally controllable currentsources, therefore requiring that servo controller 90 be configured toproperly control such current sources.

Current-mode signals 708 and 758 constitute complementary servofeedback. Feedback signal 708 is conveyed directly to inverting input107 of amplifier 100, while feedback signal 758 is conveyed directly toinverting input 157 of amplifier 150. Controllable circuitry may also beprovided to adjust the output impedance characteristics of each currentsource, such that the common-mode gain of interconnected amplifiers 100and 151 is substantially equal to the optimum value of unity.

In operation, the servo controller will continually adjust controlsignals 91 and 92 until the servo feedback currents reach a sufficientmagnitude so that main output 601 assumes an average value equal topotential 501 (typically, ground). Due to the intrinsically high inputimpedances of amplifier 100 and 150, all of the servo feedback currentwill flow only through resistors 104, 135 and 154, thereby developingvoltage drops across each resistor. At equilibrium, the average valuesat output node 601 is zero, hence the average value of the differentialvoltage present between node 108 and 158 must be zero as well, whichimplies that the DC voltage appearing across resistor 135 will be equalto the DC bias of the device connected to inputs 101 and 151.

To ensure proper operation in practical applications wherein the DUT hasno electrical connection to the amplifier circuitry other than throughinput connections 101 and 151, circuitry may be provided to supply biascurrent to the non-inverting inputs of amplifiers 100 and 150. This maybe accomplished by connecting a resistor between non-inverting input 106and reference potential 501, and connecting a substantially equalresistor between input 156 and the reference potential. To furtherimprove accuracy, the circuit can be augmented to include additionalcircuitry means to compensate for the input bias currents of each inputamplifier, such that an extremely large differentia input impedance ispresented to an external DUT by the amplifier system. Additionally,circuitry may be provided to adjust the residual input voltage offseterrors of each of the input amplifiers.

In contrast to previously described embodiments, the circuit designdepicted in FIG. 8 incorporates a generalized servo controller blockrather than just a servo integrator circuit. The controller block maycomprise a plurality of interconnected functional units, some of whichare described in detail below. Circuitry is also provided within servocontroller 90 for bi-directional communication with an external devicesuch as, for example, a microcontroller or digital signal processor. Thecommunications pass through port 94 and may include both data/controlcommands sent to the servo controller, as well as responses sent back bythe servo controller.

When the integrator within the servo controller is allowed to operatecontinuously, the low-frequency servo feedback imparts a high-passcharacteristic to the transfer function of the amplifier apparatus. Morespecifically, the nominal differential gain of the cross-coupled stageremains essentially constant for signals above the cutoff frequency, androlls off toward zero as the frequency approaches DC. For certainspecialized applications, however, is it not permissible to attenuateany low frequency signals at all while measuring the response of a DUTto an externally applied excitation, and yet the DC bias of that devicemust nevertheless be substantially blocked.

Accurate response for frequencies below 1 Hertz is particularlyimportant when measuring the time-varying potentials present onelectrochemical cells. In the case where an externally appliedexcitation current comprises square or rectangular waveforms, accuratelow frequency phase response is critical to avoid distortion in thewaveform shapes appearing across the terminal of the DUT in response tothe excitation current; this type of distortion is commonly referred toas “tilt” distortion. By using a very slow integrator, phase distortioncan be minimized, however, a very slow integrator necessarily requires aconsiderable period of time to reach equilibrium, which can betroublesome when measurements must be made rapidly on different cells.

The time to reach an equilibrium state may be reduced withoutsacrificing DC accuracy if an integrator with a nominal time constant ofseveral seconds is coupled to a ‘track and hold’ circuit, so that, onceequilibrium is established during the ‘tracking’ phase, the ‘hold’ modeis asserted whereby the servo feedback signals are held at a constantvalue just sufficient to null out the average DC signal component. Inthe “hold” mode, the servo integrator is disabled, allowing 10 anychanges in the voltage present across the DUT, even including squarewaves and abrupt DC steps, to be accurately detected, amplified andpresented at the main output. In some applications, it may be desirableto re-enable the track mode to periodically reacquire the DC bias of theDUT to accommodate very slow changes in its DC bias. This conditionarises, for example, when measurements are being made on a cell that isbeing slowly charged or discharged.

When a cell or battery is connected to a battery charger, the currentsflowing through the cell/battery will often have significant ACcomponents, for example, 120 Hertz ripple. As these currents interactwith the intrinsic impedance of the cell/battery, corresponding ACvoltages will develop within the cell that add to the intrinsic opencircuit voltage of the cell, so that the total voltage between theterminal will have both a DC component (the intrinsic voltage of thecell/battery) and an AC component. The AC component represents anunwanted noise signal. In order for the inventive amplifier to achieve astable equilibrium point when connected to such a DUT, these AC voltagecomponents must not be allowed to interfere with the action of the servofeedback loop. Therefore, controllable means, such as, for example, alow-pass filter with controllable slope and cutoff frequency, may beincorporated within the servo controller to attenuate such noise signalssufficiently that the equilibrium point will be related solely to theaverage DC potential of the cell/battery.

Depending on the particular application, the characteristics of suchexternal AC signal components can vary so widely that a singlefixed-frequency low pass filter may not suffice. Consequently, the servocontroller may incorporate a number of different function unitsrepresenting controllable filtering circuitry that can be called upon asrequired. Additional function units may include, but are not limited to:means to adjust the time constant of the integrator; a median filterwith controllable characteristics; an envelope detector; a notch filterhaving controllable center frequency, notch attenuation depth and notchwidth; signal limit detection circuitry having controllable thresholds;and a means to set the servo control output signals to predeterminedvalues. Any of these function units may be implemented using analogcircuitry exclusively or using a combination of analog and digitalcircuitry. It should be clear to those skilled in the art that suchfunction units could be adapted to operate either individually or incombination. Likewise, such function units could be incorporated asdesired within any of the embodiments of the invention without departingfrom the scope of the inventive method.

In every embodiment described thus far, the current sources have beenbipolar types capable of both sinking and sourcing current, therebyaccommodating connection to a DUT having a DC bias of either relativepolarity. Specifically, the servo control loop will be able to adjustthe feedback to provide an average DC level of zero volts (e.g., thepreferred reference potential) at the main output node 601, irrespectiveof the polarity of the DC voltage presented across inputs 101 and 151.In the special case where the DUT will always be connected with apredetermined polarity, unipolar current sources may be used as shown inFIG. 9, operative as follows.

A differential input signal is impressed between +Input 101 and −Input151, buffered by unity gain amplifiers 100 and 150, and conveyedtherefrom to amplifier 200, which by means of the connections ofresistors 201-204, converts the differential signal into a single endedoutput at output 601. Servo controller 90 comprises at least an analogor digital servo integrator. A copy of the output signal 601 is conveyedto servo controller 90, and output control signals 91 and 92 are coupledto a pair of controllable current sources each comprising an operationalamplifier, a current-sensing resistor and a transistor operative as passelement. The potential provided at non-inverting input 907 of amplifier906 is forced to appear at one terminal of resistor 905, the otherterminal of which is connected to reference ground 501, thereby causinga proportional current to flow through pass transistor 904. A secondcontrolled current source comprising elements 913, 914 and 916 operatesin the same manner, and connects to current mirror transistor 912.Thereafter, collector current from current source transistor 904 isconveyed to a current mirror comprising transistors 902 and 903, and thecollector current from current source transistor 914 is conveyed to acurrent mirror comprising transistors 912 and 913. The output currents708 and 758 of transistors 902 and 912, respectively, constitutecomplementary current-mode servo feedback. In this example, simplebipolar transistor current mirrors are used, but, without departing fromthe intended scope of the invention, Mosfet devices could be employedinstead. Likewise, other unipolar controlled current sources/topologiesand other current mirror embodiments may be substituted.

While the circuit depicted in this figure requires that the positiveterminal of a device-under-test always be connected to input 101 whileits negative terminal is connected to input 151, this convention couldbe reversed simply by interchanging the connections of the currentsource outputs 708 and 758, and reversing the relative polarity of thecontrol signals issuing from servo controller 90.

For those applications where servo feedback is needed, but truedifferential inputs are not, the differential embodiments describedabove may be reduced to their single-ended derivatives. Each of thesesingle-ended circuits operates in the same manner as its differentialcounterpart, so no additional descriptions need be provided.

FIG. 10 represents a single-ended version of the embodiment presented inFIG. 5. Here, current source 750 has been omitted entirely, and resistor203 no longer serves as an input coupling element, but is insteadconnected to reference potential 501. The servo loop will operate toestablish an equilibrium condition, irrespective of the relativepolarity of a bias potential present by a DUT connected between input101 and reference potential 501.

FIG. 11 represents a single-ended version of the embodiment presented inFIG. 6. In this variation, both input buffer 150 and current source 750have been omitted, while resistor 203 connects between node 207 andreference potential 501. The servo loop will operate to establish anequilibrium condition, irrespective of the relative polarity of a biaspotential present by a DUT connected between input 101 and referencepotential 501.

FIG. 12 represents a single-ended version of the embodiment presented inFIG. 8, wherein elements 150, 154 and 750 are all omitted, and resistor203 connects between node 207 and a reference potential 501. Inaddition, resistor 135 connects directly between node 107 and referencepotential 501. The servo loop will operate to establish an equilibriumcondition, irrespective of the relative polarity of a bias potentialpresent by a DUT connected between input 101 and reference potential501.

In an exemplary application of the present invention, small signals aremeasured at the terminals of an electrochemical accumulator. Theopen-circuit voltage of a lead-acid cell will be around 2 volts. Thecurrent-mode excitation develops small time-varying polarization voltagesignals having a peak-to-peak amplitude of perhaps 20 millivolts orless. To achieve the necessary measurement precision for these smallsignal components, amplification by at least a factor of 100 isdesirable. However, if this degree of amplification is applied directlyto the raw signal, the overall amplitude at the amplifier's output wouldreach 200 volts. Such a large output excursion cannot be readilyachieved by small signal amplifiers. Furthermore, digitization forsubsequent digital signal processing would clearly be problematic.Accordingly, a means must be employed to offset the comparatively largeDC signal component prior to amplification.

In the particular case wherein the current-mode excitation comprisessquare or rectangular waveforms, the offset compensation scheme mustallow the DC and very low frequency components of the signal of interestto pass through the system unimpeded, while still removing the large,static DC voltage of the cell/battery. This seemingly paradoxicalrequirement may be met by using a two-step process. The necessaryoffsetting compensation signal is first developed by allowing the servoloop to run to establish an equilibrium wherein the DC component hasbeen nulled out, and thereafter, the compensation is held constant atits equilibrium value. By this means, all changes in the cell's terminalvoltage occurring after the compensation has been made constant will beaccurately detected, irrespective of their spectral characteristics.

Several embodiments of the present invention have been described, and itshould be understood that variations might be made without departingfrom the scope and spirit of the present invention.

1. An amplifier apparatus comprising: differential inputs disposed toreceive a first and second input signal; a primary differentialamplifier having inverting and non-inverting input nodes receiving saiddifferential inputs through respective first and second impedanceelements and providing a main output signal; and feedback circuitrydisposed to sample said main output signal and convey complementarycurrent-mode negative feedback signals to said inverting andnon-inverting input nodes.
 2. The amplifier apparatus of claim 1,wherein said current-mode feedback signals are each servo feedbacksignals having the same low-pass characteristic.
 3. The amplifierapparatus according to claim 1 further comprising controllable circuitryto adjust common-mode gain of said primary differential amplifier at DCand at selected AC frequencies.
 4. The amplifier apparatus according toclaim 1 further comprising controllable circuitry to adjust thedifferential mode gain of said primary differential amplifier at DC andat selected AC frequencies.
 5. The amplifier apparatus according toclaim 1 further comprising controllable circuitry to adjust the inputoffset voltage of said primary differential amplifier.
 6. The amplifierapparatus according to claim 1, further comprising a unity-gain inputbuffer connected in series with each of said differential inputs.
 7. Theamplifier apparatus according to claim 6, further comprising: a biasingelement disposed between the input of the one of said unity-gain buffersand a reference potential; and a biasing element disposed between theinput of the other of said unity-gain buffers and said referencepotential.
 8. The amplifier apparatus according to claim 6 furthercomprising controllable means to adjust input offset voltages of saidunity-gain input buffers.
 9. The amplifier apparatus according to claim6 further comprising controllable means to compensate for bias currentrequired for proper operation of the non-inverting input of each saidunity-gain input buffers.
 10. An amplifier apparatus comprising: a firstinput differential amplifier disposed to receive a first input signal atits non-inverting input and output a first intermediate output signal; asecond input differential amplifier disposed to receive a second inputsignal at its non-inverting input and output a second intermediateoutput signal; a primary differential amplifier disposed to receive atits non-inverting input said first intermediate output signal conveyedthrough a first impedance element and to receive at its inverting inputsaid second intermediate output signal conveyed through a secondimpedance element, and provide a main output signal at its output node;and feedback circuitry disposed to sample said main output signal andconvey a first negative feedback signal to said inverting input of saidfirst input differential amplifier, and a second negative feedbacksignal to said inverting input of said second input differentialamplifier.
 11. The amplifier apparatus according to claim 10 furthercomprising: a third impedance element connected between the invertinginput of said primary differential amplifier and said output node ofsaid primary differential amplifier; and a forth impedance elementconnected between the non-inverting input of said primary differentialamplifier and a reference potential.
 12. The amplifier apparatus ofclaim 10 comprising: a fifth impedance element connected between theinverting input of said first input differential amplifier and theoutput of said first input differential amplifier; a sixth impedanceelement connected between the inverting input of said second inputdifferential amplifier and the output of said second input differentialamplifier; and a seventh impedance element connected between saidinverting inputs of said first and second input differential amplifiers.13. The amplifier apparatus according to claim 10, further comprising: abiasing element disposed between the non-inverting input of a firstinput differential amplifier and a reference potential; and a biasingelement disposed between the non-inverting input of a second inputdifferential amplifier and a reference potential.
 14. The amplifierapparatus according to claim 10 further comprising circuitry to adjustcommon-mode gain of at least one of said input differential amplifiersat selected frequencies.
 15. The amplifier apparatus according to claim10 further comprising circuitry to adjust differential-mode gain of atleast one of said input differential amplifiers at selected frequencies.16. The amplifier apparatus according to claim 10 further comprisingcircuitry to compensate for bias current required for proper operationof the non-inverting input of each said differential amplifier.
 17. Theamplifier apparatus according to claim 1 wherein said complementarynegative feedback signals have substantially identical magnitudes. 18.The amplifier apparatus according to claim 1 wherein said feedbackcircuitry comprises: an integrator; and a first and second controllablecurrent-source controlled by said integrator, said current sourcesproviding current-mode output signals of opposite relative polarity. 19.The amplifier apparatus according to claim 18 wherein said integratorfurther comprises means to produce an inverted signal controlling atleast one said current source.
 20. The amplifier apparatus according toclaim 18 wherein said integrator comprises: an analog-to-digitalconverter disposed to sample said main output signal and provide adigital output signal; a virtual integrator receiving said digitaloutput signal and providing a digital integral signal representing thetime-integral said main output signal; and a digital-to-analog converterthat receives said digital integral signal and provides a correspondinganalog signal representing a control signal adapted to control saidfirst and second current sources.
 21. The amplifier apparatus accordingto claim 18 wherein: said first controlled current source provides afirst bipolar output current that is inversely proportional to a controlsignal from said integrator; and said second controlled current sourceprovides a second bipolar output current that is directly proportionalto said control signal.
 22. The amplifier apparatus according to claim18 wherein: said first controlled current source provides a firstbipolar output current that is inversely proportional to a controlsignal from said integrator; and said second controlled current sourceprovides a second bipolar output current that is directly proportionalto said control signal.
 23. The amplifier apparatus according to claim18 wherein said first controlled current source provides a firstunipolar output current that is inversely proportional to a controlsignal from said integrator; and said second controlled current sourceprovides a unipolar output current that is directly proportional to saidcontrol signal from said integrator.
 24. The amplifier apparatusaccording to claim 1 further comprising: a servo controller; andcircuitry adapted to communicate with an external controller.
 25. Theamplifier apparatus of claim 1 further comprising: track and holdcircuitry coupled to said feedback circuitry.
 26. The amplifierapparatus according to claim 18 further comprising: a controllablelow-pass filter disposed to reduce the high frequency content of signalsinput to said integrator.
 27. The amplifier apparatus according to claim18 further comprising: a controllable notch filter disposed to reducethe amplitude of a predetermined range of frequencies within signalsinput to said integrator.
 28. The amplifier apparatus according to claim18 further comprising: a controllable impulse noise discriminatordisposed to reduce the impulse noise content of signals input to theintegrator.
 29. The amplifier apparatus according to claim 1 furthercomprising circuitry for developing a control signal for at least onecontrolled voltage source.
 30. A signal detection and amplificationsystem comprising amplifier apparatus having: differential inputsdisposed to receive a first and second input signal; a primarydifferential amplifier receiving said differential inputs and providinga main output signal; and feedback circuitry disposed to sample saidmain output signal and convey complementary current-mode negativefeedback signals to said differential inputs.
 31. The signal detectionand amplification of claim 29 adapted to test electrochemical cells. 32.The signal detection and amplification system of claim 29 adapted tomonitor a plurality of electrochemical cells.
 33. The signal detectionand amplification system of claim 29 adapted to provide feedback to acharger for electrochemical cells.
 34. The signal detection andamplification system of claim 29 adapted to analyze impedance ofelectrochemical cells.
 35. A method for compensating the output of anamplifier system comprising: integrating said output to generate afeedback control signal; controlling a pair of complementary currentmode feedback signals with said feedback control signal; connecting saidpair of complementary current mode feedback signals to input nodes of aprimary amplifier wherein said input nodes are connected throughimpedance elements to differential inputs of said amplifier system. 36.The method according to claim 35 wherein said feedback signal causessaid amplifier system to exhibit a high pass characteristic andattenuate a predetermined low-frequency component of said output. 37.The method according to claim 35 comprising: coupling a differentialinput signal to inputs of each of two unity gain amplifiers, whereby thefirst unity-gain amplifier produces a first intermediate output signaland the second unity-gain amplifier produces a second intermediateoutput signal; conveying said first intermediate output signal through afirst impedance element to the non-inverting input of a primarydifferential amplifier; and conveying said second intermediate outputsignal through a second impedance element to the inverting input of saidprimary differential amplifier that provides a main output signal. 38.The method according to claim 35 wherein said integrating step isperformed by a controllable integrator including only analog circuitry.39. The method according to claim 35 wherein said step of integrating isperformed by a controllable integrator including a combination of analogand digital circuitry.
 40. An amplifier apparatus comprising: a primaryamplifier providing a main output signal; and feedback means disposed tosample said main output signal and convey complementary current-modenegative feedback signals to said primary amplifier.
 41. An amplifierapparatus according to claim 40, wherein the DC value of said mainoutput signal is substantially equal to a reference potential.